Transmitter, transmission method and receiver

ABSTRACT

A transmission method and a transmitter and receiver Structure is disclosed which provides for transmission of modulated waves using long pulses with a plurality of frequencies. The method involves two consecutive frequencies being separated by 1/T, where T is the period of the useful transmission intervals. The method and the apparatus are particularly suited to broadcasting and reception of television and radio signals as well as telephone communications between exchanges and between radio telephones and communication stations including terrestrial stations and satellites and local computer networks. Most particularly the method is applicable to high fidelity radio transmissions as well as to high definition television (HDTV).

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates to a transmitter as well as a transmission methodwhich are particularly high-performance. The invention also relatesprincipally to a receiver.

2. Discussion of Background

It is known to transmit ion by using modulated waves, such as, forexample, electromagnetic waves. It is known to try to increase thetransmitted information throughputs. However, the increase in thethroughput more often than not comes through an increase in the band offrequencies used. Now, in the case of guided transmissions there is alimitation due to the passband of the channel, for example a coaxialcable or an optical fibre, whilst in transmissions over the air thereare no longer sufficient frequencies to satisfy all the needs.

Moreover, the Patent Application FR 86 09622 published under the No.2,601,210 describes a method of transmission using symbols A (f, t) .Each symbol corresponding to a frequency and a given transmission time.The transmission time not being fixed, the device does not use any meansof fine synchronization, which thus limits the spectral response toapproximately 0.7 bit/(Hz.s). Moreover, the French certificate ofaddition 86 13271 published under the No. 2,604,316 describes the use ofdevices for calculation of the discrete Fourier transform for thedemodulation of the signal. This certificate of addition suggestmoreover the use of a guard period between the symbols. However, thenonorthogonality of the channels used limits the spectral response, inthe best of cases, to 1 bit/(Hz.s).

With the device according to the present invention, it is possible toexceed 5 bits/(Hz.s) under analogous conditions.

The present invention relates to an improvement in the means and oftransmission methods described in the Patent 86 13937; 86 13938; 8613939; 86 13940; 86 13941; 86 18351; 86 18352.

In the devices of known type it has often been tried to increase theinformation throughput by decreasing the transmission times allocated toeach information unit, (or by increasing the number of possiblesymbols). Thus, a broadened spectrum was generated, whose secondarylobes have to be filtered on transmission, thus creating a distortion inthe signal. For a square signal with a period τ a spectrum possessingnumerous secondary lobes is obtained; the principle has a width of 2/τ.In the remainder of this specification this distortion in the signal iscalled auto-distortion.

SUMMARY OF THE INVENTION

The device according to the present invention offers the originalfeature of reducing or eliminating the auto-distortion of the signal byusing long transmission intervals for the information elements (oftencalled symbols) to be transmitted. To obtain a high throughput aplurality of information elements are simultaneously transmitted byusing orthogonal channels. An information element is, for example, adigital word of 6 bits. Advantageously, one information element pertransmission channel is transmitted. The orthogonality at the receivingend of the transmission channels enables the separation of theinformation belonging to various channels. The orthogonalization at thereceiving end of the various channels is the result of a choice ofseveral transmission frequencies regularly spaced by k/T, k is a naturalnumber and T the period of the useful transmission interval. This typeof transmission assumes a synchronous sampling at the receiving end inorder to obtain the separation of the channels.

The subject of the invention is principally a method for transmission ofmodulated waves simultaneously using a plurality of frequencies,characterized in that it comprises successive steps of transmission ofdigital words for a period T+ΔT, two transmission frequencies beingseparated by 1/T, T being the useful transmission interval and ΔT beingthe transition interval.

The subject of the invention is also a method characterized in thatΔT>0.

The subject of the invention is also a method characterized in thatsynchronization signals are transmitted enabling, at the receiving end,the sampling of the signal for useful transmission intervals of period Tso as to render orthogonal channels corresponding to the variousfrequencies.

The subject of the invention is also a method characterized in that T islarge relative to ΔT.

The subject of the invention is also a method lo characterized in thatthe first frequency used for is equal to k/2T, k being a positiveinteger or zero.

The subject of the invention is also a method characterized in that thetransmission is stopped during the transition intervals.

The subject of the invention is also a method characterized in that itcomprises a step:

for determination of the patterns for the useful transmission intervalof period T,

of transmission of the pattern for a transmission interval of period Tand its coherent continuation during the transition interval of periodΔT.

The subject of the invention is also a method characterized in thatduring each useful transmission interval of period T a digital word istransmitted on each frequency.

The subject of the invention is also a method characterized in thatduring each transmission interval of period T a (real part, imaginarypart) or (amplitude, phase) pair is transmitted on each frequency, the(real part, imaginary part) or (amplitude, phase) pair being inone-to-one equivalence with the information to be transmitted.

The subject of the invention is further a transmitter characterized inthat it enables the implementation of the method.

The subject of the invention is also a transmitter characterized in thatit comprises a modulation device enabling the transmission, during auseful transmission interval of period T, of a digital word on eachfrequency used.

The subject of the invention is also a transmitter characterized in thatthe modulation device comprises N modulators, N being the number offrequencies used; the outputs of the N modulators being connected to theinputs of a summation device.

The subject of the invention is also a transmitter characterized in thatthe summation device comprises a symmetrical distribution tree.

The subject of the invention is also a transmitter characterized in thatthe modulation device comprises a device for calculation of the inverseFourier transform.

The subject of the invention is also a transmitter characterized in thatthe device for calculation of the inverse Fourier transform is a digitalcircuit for calculation of the fast Fourier transform (FFT).

The subject of the invention is also a transmitter characterized in thatone of the transmission channels is centred on the zero frequencycarrier.

The subject of the invention is also a transmitter characterized in thatthe modulation device operates at intermediate frequency.

The subject of the invention is also a transmitter characterized in thatthe modulation device is a digital device for carrier modulation.

The subject of the invention is also a transmitter characterized in thatit comprises means of generation, on at least some of the frequenciesused, of calibration signals for the amplitude A and/or for the phase φ.

The subject of the invention is also a transmitter characterized in thatthe said transmitter is a transmitter of digital data.

The subject of the invention is also a transmitter characterized in thatthe said transmitter is a television transmission transmitter.

The subject of the invention is also a transmitter characterized in thatthe said transmitter is a radio transmission transmitter.

The subject of the invention is also a receiver comprising means forsampling, synchronous with the signal, characterized in that itcomprises means for demodulation of a modulated wave transmission usingsymbols transmitted for a period T+ΔT on a plurality of frequencies, twotransmission frequencies being separated by 1/T, T being the usefultransmission interval and &T being the transition interval, and that itcomprises a servocontrol device ensuring the synchronization of thereceiver with the received signal.

The subject of the invention is also a receiver characterized in that itcomprises an automatic gain control device (AGC) controlled by a devicefor detecting the mean power of at least part of the signal.

The subject of the invention is also a receiver characterized in that itcomprises (real part, imaginary part) or (amplitude, phase) pairdecoding means, in order to convert them into digital words.

The subject of the invention is also a receiver characterized in that itcomprises at least one device for calculation of the fast Fouriertransform (FFT).

The subject of the invention is also a receiver characterized in that itcomprises a test device capable of supplying reference phases and/oramplitudes from calibration signals.

The subject of the invention is also a receiver characterized in that itcomprises a device for equalization, compensating for the perturbationsin the signal coming from the transmission.

The subject of the invention is also a receiver characterized in that itcomprises reorthogonalization means using the transition interval ofperiod ΔT in order to render a plurality of channels orthogonal.

The subject of the invention is also a receiver characterized in thatthe said receiver is a receiver of radiophonic transmissions.

The subject of the invention is also a receiver characterized in thatsaid receiver is a receiver of television transmissions.

The subject of the invention is also a method characterized in that themodulated waves are electromagnetic waves.

The subject of the invention is also a method characterized in that theN orthogonal channel separation step comprises a step for calculation ofthe fast Fourier transform (FFT) of the signal.

The subject of the invention is also a method characterized in that itcomprises a step for reconstruction of a television signal from signalsreceived in the N channels.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be better understood by means of the descriptionhereafter and the attached figures given as nonlimit examples, in which:

FIG. 1 is a diagram illustrating the spectrum broadening phenomenon;

FIGS. 2a and 2b is a diagram illustrating the transmission of a carrierfrequency;

FIG. 3 is a diagram explaining the operating principle of the deviceaccording to the present invention;

FIG. 4 is a diagram explaining the operating principle of the deviceaccording to the present invention;

FIG. 5 is a diagram explaining the operating principle of the deviceaccording to the present invention;

FIGS. 6a-6c comprises chronograms illustrating the chaining of theperiods or transmission intervals;

FIG. 7 is a diagram of an example of encoding capable of beingimplemented in the device according to the present invention;

FIG. 8 is a general diagram of a transmitter according to the presentinvention;

FIG. 9 is a diagram of a first embodiment of a transmitter according tothe present invention;

FIG. 10 is a diagram of a second embodiment of a transmitter accordingto the present invention;

FIG. 11 is a diagram of a third embodiment of a transmitter according tothe present invention;

FIG. 12 is a diagram of an embodiment of a detail of the transmitters ofFIGS. 9, 10 or 11;

FIG. 13 is a diagram of an embodiment detail of the transmitters in FIG.11;

FIG. 14 is a diagram of a first embodiment of a detail of thetransmitters according to the present invention;

FIG. 15 is a diagram of a second embodiment of a detail of thetransmitters according to the present invention;

FIG. 16 is a diagram of a first embodiment of a detail of the deviceillustrated in FIG. 13;

FIG. 17 is a diagram of a second embodiment of the device illustrated inFIG. 13;

FIG. 18 is a curve illustrating the information throughput obtained as afunction of the coding state number for a period of the usefultransmission interval T and a number of given channels used;

FIG. 19 is a curve illustrating an analog embodiment of thetransmitter-receiver synchronization.

FIG. 20 is a diagram of a third embodiment of a detail of thetransmitters according to the present invention;

FIG. 21 is a diagram of an embodiment of a receiver according to thepresent invention;

FIG. 22 is a diagram of an embodiment of a television receiver accordingto the present invention;

FIG. 23 is a diagram illustrating an example of equalization capable ofbeing implemented in the device according to the present invention;

FIG. 24 is a diagram of an architecture capable of being implemented inthe device according to the present invention;

FIG. 25 is a diagram of an embodiment of a detail of the receiveraccording to the present invention;

FIG. 26 is a curve illustrating an analog embodiment of thetransmitter-receiver synchronization;

FIG. 27 is an embodiment of a device capable of being implemented in thedevice according to the present invention;

FIG. 28 a curve illustrating the information throughput obtained as afunction of the coding state number for a period of the usefultransmission interval T and the number of given channels used.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

In FIG. 1 can be seen a curve 3 representing, at the receiving end, theamplitude A 2 of the spectrum of a constant amplitude wave transmittedfor a limited interval of time of period T. The curve 3 has the form ofsin x/x.

In frequency, besides a principal lobe, secondary lobes are transmittedwhich continue to diminish in step with the distancing from the centralfrequency f₀. The amplitude A passes through zero at two points,referenced 50 and 51, symmetrical relative to the frequency f₀. Thepassages through the zero amplitude are regularly distributed, separatedby 1/T.

The broadening of the spectrum depends principally on the period of thetransmitted pulses. The short transmissions cause a greater frequencybroadening. In the devices of known type, the broadening of the spectrumwith a limited passband allocated to the transmissions, prolongs theperiod of the pulsed response of the signal, thus creating interferencesbetween the pulses (called "intersymbol interferences"). The quantity ofseparable information was thus limited.

In FIG. 2a can be seen a curve 6 corresponding to the transmission of apure sinusoidal frequency starting from an instant 16. The signal 6 can,for example, correspond to a carrier. The curve 6 shows the amplitude ksa function of time.

In FIG. 2b can be seen a curve 7 showing the amplitude, as a function oftime, of the wave 6 received by a receiver. Insofar as the receiver isfixed relative to the transmitter, the received wave 7 has the samefrequency as the transmitted wave 6. However, the amplitude and thephase have changed. In FIG. 2b, the start of the reception carries thereference 17. The instant 17 is later than the instant 16, thedifference corresponding to the propagation time of the waves betweenthe transmitter and the receiver. Starting from an instant 18, thesignal 7 has the same shape as the signal 6. Between the instant 17 andthe instant 18 we witness the establishment of the signal during whichvarious perturbations are observed. The perturbations in the timeinterval between the instant 17 and the instant 18 are principally theresult of distortions brought about by the transmission and receivingequipment, perturbations due to multiple echoes, as well as to thelimitation of the passband of the transmitter. Insofar as the frequencyof the signal 6 and of the signal 7 is conserved, it is possible, bycarrying out a calibration of the amplitude and of the phase, to regain,at the receiving end, the information transmitted. Certain frequencymodifications, as for example the frequency modification due to theDoppler effect owing, for example, to the displacement of the receiverrelative to the transmitter, will be able to be compensated by asuitable calibration.

In order to carry out the calibration it is considered, for example,that everything that occurs between the transmitter and the receiver isa filter having a stable frequency response over periods much largerthan the period T of the symbols. By transmitting signals of known type,it is possible to determine the frequency response of the filter. Thus,by applying the inverse frequency response the regeneration, at thereceiving end, of the transmitted signal is accomplished.

In FIG. 3 can be seen a diagram illustrating the operating principle ofthe device according to the present invention. In FIG. 3 can be seen afirst curve 31 and a second curve 32 which are centred on frequencies f₀and f₀ +1/T respectively, T being the period of a useful transmissioninterval. The amplitude A of the curve 31 passes through zero at thepoints 51 and 53.

The amplitude A of the curve 32 passes through zero at the points 52,54.

The point 52 corresponds to the maximum amplitude of the curve 31 and tothe zero amplitude of the curve 32. A point 311 of the spectrum at thefrequency f₀ is not perturbed by the signal corresponding to the curve32.

In the same way, the point 53 corresponds to the maximum amplitude ofthe curve 32 and to the zero amplitude of the curve 31. At the point 53,at the frequency f₀ +1/T the signal belongs solely to the curve 32. Bysampling the spectrum at the frequencies f₀ and f₀ +1/T the completeseparation of the frequencies corresponding to the curves 31 and 32 isaccomplished. It will readily be possible to use independent amplitude,phase or amplitude/phase states on each of the frequencies f₀ and f₀+1/T. The two codings being perfectly independent and separable at thereceiving end, it is possible to distribute the total informationthroughput between several channels.

In the devices of known type, in order to increase the transmittedinformation throughput the period of the pulses and/or the time reservedfor the transmission of each elementary item of information wasdecreased (or by increasing the number of possible symbols).

In contrast, in the device according to the present invention for agiven information throughput, insofar as it is possible to distributethe throughput to be transmitted between several channels, it can beallowable to increase the period T of the pulses and/or of the usefultransmission intervals corresponding to an elementary item ofinformation. The total throughput being obtained by effecting the sum ofthe elementary throughputs corresponding to each frequency. Byincreasing the period T of the useful transmission intervals thebroadening of the spectrum and the auto-distortion of the signal aredecreased. It it thus possible, as illustrated in FIG. 4, to use a largenumber of carriers 31 to 3N. The use of N frequencies 31, 3N enables avery extensive filling of the passband B. As in the case of FIG. 3,successive curves are separated in frequency, by 1/T. Thus, the maximumin the spectrum of each channel corresponds to the passage through thezero amplitude of the spectra of all the other channels.

In FIG. 4, a curve 3i passes through the maximum at a frequencycorresponding to the point 5 (i+1) and through the zero amplitude at thefrequencies corresponding to the points 5j j≠i+1. For clarity in theFIG. 4, only the secondary lobes of the curve 31 have been shown.

Each transmission channel corresponding to a different carrier frequencytransports an item of information independent from the other channels.The total throughput is equal to the sum of the throughputs of Nchannels.

The increasing of the number of channels increases the period T of theuseful transmission intervals, without decreasing the throughput.

In contrast, the increasing of the number of channels necessitates, atthe transmission and at the reception ends, larger or higher-performancehardware.

For a correct operation of the device according to the presentinvention, it is imperative that the useful part of the signal, at thereceiving end, be stable. To achieve this, at the receiving end, thetime intervals of period ΔT, during which the signal risks not beingstationary, are eliminated. This time corresponds principally to thepulsed responses of the transmitter and of the receiver and to themultiple paths. In the remainder of this Patent, the interval duringwhich the signal risks not being stationary is called transitioninterval of period ΔT, the stationary part of the signal exploited bythe receiver, useful interval of period T. The signal is advantageouslytransmitted for transmission intervals of period T+ΔT. Thus, the spectraof each channel although uniformly distributed with spacing of 1/T, havea width of the principal lobe equal to 2/(T+ΔT). At the receiving end,only the useful interval of period T is used, which thus enablesreconstruction of the spectrum of FIG. 4. A nonlimiting example ofthroughput in M bits/s is shown in FIG. 18 as a function of the numberof possible states (that is to say of different symbols capable of beingtransmitted).

Furthermore, the number of bits of the coding has been indicated asabscissa. For example, a coding on 4 bits gives 2⁴ =16 different states.The curves are given for an identical transition interval of period ΔT=8μs.

A first curve indicates the throughputs obtained for N=64 and T+ΔT 16μs.

A second curve indicates the throughputs obtained for N=128 and T+ΔT=24μs.

A third curve indicates the throughputs obtained for N=256 and T+ΔT=40μs.

A fourth curve indicates the throughputs obtained for N=512 and T+ΔT=72μs.

For constant transition interval, the fact of increasing the useful partof the signal leads to a saturation phenomenon limiting the throughout,not shown in FIG. 18. The curves in FIG. 18 correspond to a passband Bof 8 MHz.

According, in particular, to the available passband, the application andthe propagation conditions, those skilled in the art will choose theideal compromise between the number of channels N and the usefultransmission interval T.

The throughput can be increased up to a certain limit, by using atransition interval of very small period ΔT relative to the period T ofthe useful transmission interval.

It is advantageous to use the inverse fast Fourier transform (FFT⁻¹) tocarry out the modulation of the channels at the transmitting end and thefast Fourier transform (FFT) to carry out the demodulation at thereceiving end. The use of fast Fourier transform algorithm imposes thecarrying out of the calculations on a number of samples equal to a powerof two. In the course of television transmissions 256, 512, 1024 or 2048channels are, for example, used. However, it is not necessary for eachchannel to transmit an item of information.

At the receiving end, for each useful transmission interval T the phaseand the amplitude corresponding to each of the frequencies 31 to 3N ismeasured advantageously. A synchronous sampling is used to extract theitem of information from the signal.

The amplitude representing the item of information is constant over allthe period of the transmission interval of period T or T+ΔT and thephase representing the item of information corresponds to the phaseshifting relative to a phase reference.

A receiver dedicated to the reception of the waves transmitted by thetransmitter according to the present invention is described in a FrenchPatent Application filed by the Applicant simultaneously with thepresent Patent Application and bearing the immediately higher number.

In order to obtain a large information throughput it is necessary to beable to distinguish close phases and amplitudes and hence to make use ofa phase and amplitude reference for each channel. This reference for theamplitude and for the phase is advantageously given by reference signalsperiodically transmitted by the transmitter towards the receiver. Thefrequency of repetition of the reference signals depends on thestability of the propagation conditions and the local oscillators.

In a first variant of the device according to the present invention,phase and amplitude reference signals are periodically transmitted onall the frequencies 31 to 3N of a time interval of period T or T+ΔT.However, it is necessary to note that the frequent transmission ofcalibration signals reduces the throughput of useful transmittedinformation.

In an advantageous variant of the device according to the presentinvention only a few calibration signals are transmitted regularlydistributed advantageously over the frequencies 31 to 3 N, thecoefficients of the other frequencies being determined by calculation,for example by interpolation.

More generally, it is possible to distribute the calibration signals intime and/or on different channels.

It is for example possible to periodically transmit test signals, eachtransmission being made on different channels. For example, a circularpermutation of the channels assigned to the tests is carried out. Thepulsed response of the transmission medium is deduced for all thechannels, for example, by interpolation in time and/or on thefrequencies. The matrix of the corrections in amplitude and in phase tobe applied to each channel is thus deduced.

It is paramount to compensate, by calibrations, the variations in thepulsed response of the transmission medium, for example owing to avariation (even local) in the atmospheric conditions.

The pulsed response of the medium is determined, for example, bycalculating the Fourier transform of the corrections to be applied.

In one embodiment a channel at the 8 level served in the calibration ofthe amplitude A and of the phase of all the channels 31 to 3N. In such atype of device it is possible, either to carry out the calibration ateach useful transmission interval of period T, or as in the case of apreviously described embodiment, to solely reserve certain transmissionintervals for the calibration. The synchronization is maintained by theuse of a stable time base.

The number of channels and/or of periods reserved for the calibrationdepends on the error that it is wished to be able to correct, as well ason the perturbations which are capable of affecting the informationtransmission. The calibrations will, for example, have to be morefrequent, to compensate the displacement of the frequencies by Dopplereffect in the case of displacements of one transmitter relative to theother, for example, in the case of radiotelephones or of communicationsbetween aircraft.

The first and the last channels risk being perturbed, in particular, bythe filters of the transmitter and of the receiver. Advantageously asillustrated in FIG. 5, the first and the last channel are not used forthe information transmission. For example, nothing is transmitted on thefirst and the last channel or the transmission of the second channel isrepeated on the first and of the penultimate on the last.

In FIG. 6 can be seen various examples of chronograms for chaining thesuccessive transmission periods 8.

In FIG. 6a can be seen useful transmission intervals 8. Between theuseful transmission intervals 8 there are transition intervals 81 notoffering any decrease in transmitted power. For example, the signaltransmitted at the end of the corresponding useful transmission intervalis transmitted in the transition intervals 81. The fact that thetransmission power is not decreased enables the best use of theamplifiers of the transmitters.

In FIG. 6b can be seen successive useful transmission intervals 8 whichare not separated by transition intervals. This case corresponds to themaximum information throughput. It has the disadvantage of the lowsecurity of transmission in the event of perturbations. This variantwill for example be used for the cable transmissions.

In FIG. 6c can be seen a succession of useful transmission intervals 8separated by transition intervals 81 during which the transmission ofmodulated waves is stopped. There is thus a saving in energy.

The choice of the type and of the period of transmission intervals 81depends on the hardware used and on the hoped-for transmission andreceiving conditions. For example, if large multiple echoes areexpected, the use of longer transition intervals will be well advised.The length of the transition interval 81 will for example be determinedfrom worse conditions in which it is desired to be certain of obtaininga correct reception. For example, if it is desired to be able to be freefrom multiple echoes coming from a maximum distance of 600 meters, atransition interval 81 will be used corresponding to the propagationtime of this echo, for example electromagnetic, and possibly of the timecorresponding to the damping of the pulsed response of this echo, forexample 4 μs.

In FIG. 7 can be seen an example of information coding capable of beingimplemented in a device according to the present invention. This type ofcoding has been described in the Patent Applications FR 86 13937; FR 8613938; FR 86 13939; FR 86 13940; FR 86 13941; FR 86 18351; FR 86 18352.In this type of coding an amplitude and a phase in the complex plane isassociated with each digital word. The (amplitude, phase) pair isequivalent to the real and imaginary part of the signal. In the exampleillustrated the (amplitude, phase) pairs 14 are regularly distributed onthe concentric circles 150, 160, 170 and 180. In the example illustratedin FIG. 7 use is made of 32 different values, which corresponds to acoding on five bits. It must be understood that the coding on adifferent number of bits, as for example 2, 3, 4 or 6 or more, does notdepart from the scope of the present invention. The size of the discs 13centred on the points 13 corresponding to the same digital word, enablesthe toleration of a certain inaccuracy. The greater is the diameter ofthe discs 13 and the lower will be the rate of errors, but the less willit be possible to have different values. In the example illustrated inFIG. 7, the circles 150, 160, 170 and 180 have diameters ρ1, ρ2, ρ3 andρ4 respectively equal to √2/2, 1, ρ2 and 2, the power of a transmitterbeing normalized to 1. In the example of FIG. 7, in order to decreasethe risks of errors at the receiving end, the discs 13 are spread to themaximum. Thus, on each subsequent circle, the points 14 are placed onthe bisector of points 14 of the previous circle. It must be understoodthat the arrangement in FIG. 7 is given only by way of nonlimitingexample. For example, the distribution of the points 14 on a rectangleor a spiral, for example logarithmic or Archimedian does not depart fromthe scope of the present invention. In the same way any other type ofcoding can be used, the type of coding depends on the throughput and onthe nature of the item of information to be transmitted. The coding canbe analog or digital according to the desired application.

In the device according to the present invention, it is possible tocarry out the analysis of the pulsed response of the transmissionmedium. According to the application it is possible to use a real timeanalysis or a deferred analysis.

The analysis enables adaptation of the transmission standard to thelocal conditions, for example in a local computer or telephone networkor in directional radio links.

For example, on a local network it is possible to carry out the analysisat each reconfiguration of the network. In order to eliminate thereflections in the cables parts of the transition interval (of totalperiod ΔT) can be placed at the moment when these reflections aregreatest.

In directional radio links a computer is for example used to carry outthe real time analysis of the pulsed response of the medium and to adaptthe transmissions in such a way as to obtain the maximum throughputpermitted by the perturbations of the medium. For example, the computerdecreases the period ΔT of the transition interval when this is possiblewithout exceeding the accepted error rate. In one variant, the computercarries out the choice of a means of transmission from amongst aplurality of available ones.

In FIG. 8 can be seen a general diagram of an embodiment of atransmitter according to the present invention. The transmittercomprises a coding device 70 and a modulation device 90.

The coding device 70 receives information to be transmitted frominformation sources 73. The information sources can be, for example, atelevision camera, a microphone, a video recorder, a tape recorder, atelevision control room, a computer, a telephone exchange, a dataacquisition device, a radiotelephone, a telephone, an information sourceassociated with a radar, a sonar and/or a sensor. Advantageously, thetransmitter according to the present invention comprises, between theinformation sources 73 and the coding device 70, a device 700 forinformation processing enabling the desired modifications to be carriedout. For example, the information processing device comprises a deviceof known type for information throughput reduction, for example byelimination of redundant information. Advantageously, the device 700comprises a device for scrambling the signal of known type whichsupplies a signal comprising the item of information to be transmitted,but whose integration in time corresponds to a white noise. Insofar as,on the one hand, the device according to the present invention enablesthe transmission of large information throughputs, and on the otherhand, it is possible to transmit, either simultaneously, or bytime-division multiplexing, different types of information, it ispossible to simultaneously connect several sources 73 to the codingdevice 70. The coding device 70 carries out the coding either to obtainthe highest performance or to conform to an established transmissionstandard. The item of processed information is transmitted from thecoding device 70 to the modulation device 90. The modulation device 90enables the simultaneous modulation of a plurality of carriers asillustrated, for example, in FIG. 4. The signals modulated by themodulation device 90 are amplified by an amplifier 77, transmitted, forexample by an aerial 40, or injected into a cable 400. If it proves tobe necessary, the modulation of a high frequency carrier is carried outbefore the transmission.

Insofar as N independent channels are transmitted, it is possible tocarry out the separate amplification of various channels.

In FIG. 9 can be seen an embodiment of transmitters according to thepresent invention comprising a plurality of amplifiers 77 positionedbetween the modulation device 90 and the summation device 76.Advantageously, each amplifier 77 corresponds to one channel. However,it is possible without departing from the scope of the presentinvention, to assign several amplifiers 77 to each channel or, on thecontrary to carry out a partial summation of several channels on outputfrom the modulation device 90 in order to apply them to a singleamplifier.

The use of a plurality of amplifiers 77 is particularly suited totransistorized amplifiers. In fact, it is known to use the sum of thepowers supplied by a plurality of transistorized modules in order toobtain the desired power.

In FIG. 10 can be seen a first embodiment of the transmitter accordingto the present invention. In the example illustrated in FIG. 10 thesignal to be transmitted is supplied by a television camera 71, amicrophone 72 and/or other sources 73. Advantageously, the sources 71,72 and/or 73 are connected to the information processing device 700. Thecoding device 70 comprises a shaping circuit connected to a device forcomplex digital/signals conversion. The modulation device 90 comprises aset of N modulators, referenced 91 to 9N, connected to a summationdevice 76. The device for summation of the signal 76 comprises, forexample, a symmetrical distribution tree 760. The modulation device 90is connected to an amplification device 77 itself connected to atransmission aerial 40 and/or a transporting cable 400. Theamplification device 77 can comprise frequency elevation devicesnecessary to satisfy the transmission standards.

The shaping device 74 produces the desired shape of the signals comingfrom the sources 71 to 73. For example, the shaping device 74 carriesout the multiplexing of the various sources and supplies numbers inseries. The shaping circuit 74 comprises sampling circuits,analog-to-digital converter circuits, and/or multiplexors. In the caseof digital devices, the calculating power of the shaping device 74depends principally on the desired information throughput. For example,a high definition digital television transmission with high fidelitystereophonic sound in several languages as well as digital informationwill demand a much greater throughput than, for example, a stereophonicradiophonic transmission, or all the more so, a radiotelephonetransmission.

Advantageously, (amplitude, phase) pairs, for example like thoseillustrated in FIG. 7, or (real part, imaginary part) pairs from thesignal are transmitted. The device for digital conversion complexsignals 75 generates, from digital words supplied by the shaping device74, (real part, imaginary part) or (amplitude, phase) pairs from thesignal and distributes them between the various modulators 91 to 9N. Thesummation device 76 supplies, to the input of the amplification device77, a composite signal comprising the frequencies 31 to 3N necessary forthe transmission. The frequencies 31 to 3N are modulation frequencies.Thus, it is possible, either at the modulation device 90 level or at theamplification device 70 level to raise the transmission frequency. Thecomposite signal carried on a, for example, high frequency carrier istransmitted by the aerial 40 or is injected into the cable 400.

In FIG. 11 can be seen a second embodiment of the transmitter accordingto the present invention. The device in FIG. 11 comprises, between theoutput of the conversion device 75 and the input of the amplifier 77, adevice for rearrangement of the signal 78, a device for calculation ofthe inverse Fourier transform 190, a device for serialization of thesignal 301 and a device for carrier signal generation 302 which areconnected in series. The modulation of the composite signal to betransmitted can be obtained by calculating an inverse Fourier transform.

Advantageously, a computer 190 capable of calculating a discrete inverseFourier transform is used.

Advantageously, an inverse fast Fourier transform (FFT⁻¹) calculationcircuit is used. The use of inverse fast Fourier transform algorithmsimplies that the number N of channels be a power of 2. However, it isnot necessary that all the channels carry information.

Demonstration of the possibility of using discrete inverse Fouriertransform algorithms to carry out the modulation of the signal:

Let N frequencies f₀, f₀ +1/T, f₀ +2/T, f₀ +3/T, . . . , f₀ +k/T, . . ., f₀ +(N-1)/T be amplitude and/or phase modulated for a time interval ofperiod T. The N modulated carriers are:

    Sk(t)=Ak exp (j(2π(f.sub.0 +k/T)t+φk))

k being an integer lying between 0 and N-1

Ak being the amplitude of the order k carrier,

t being the time

φ k being the phase of the order k carrier.

Assuming that the reference in the transmitted phase value is taken atthe start of the time intervals T.

The signals Sk (t) and Sk' (t) are independent and completely separableif they fulfil the orthogonality condition: ##EQU1##

The orthogonality condition is therefore satisfied if 4πf₀ T=2Iπ, Ibeing an integer, which is equivalent to

    f.sub.0 =I/2T.

Taking a frequency f₀ -(N/2-1)/T=(2L-N)/2T.

Sampling the signals Sk (t) at the sampling frequency f₀ =N/T=B, B beingthe passband. ##EQU2##

The modulated signal X can be written ##EQU3##

Setting k'=k+(N/2)+1 for k lying between 0 and (N/2)-2, that is to saythat k' lies between (N/2)+1 and N-1, and k'=k-(N/2)+1 for k lyingbetween N/2-1 and N-1 which corresponds to k' lying between 0 and N/2.##EQU4## {X (n))} is the discrete inverse Fourier transform {(DFT⁻¹) ofA((N/2)-1) exp (j φ (N/2)-1), . . . , AN-1 exp (j φ (N-1)), . . . , Aoexp (j φ 0), . . . , A ((N/2)-2) exp (j φ (N/2)-2)

Similarly, at the receiving end, it is possible to carry out thedemodulation of the signal by carrying out a discrete Fourier transform(DFT).

The invention is not limited to the use of the inverse Fourier transformto carry out the modulation of the signal. Other algorithms transforminga frequency domain into the time domain can be implemented.

The serialization device 301 supplies, advantageously, a succession ofdigital values to the device for generation of the signal 302.Advantageously, the serialization 301 repeats certain digital values insuch a way as to generate the transition interval. Advantageously,during the transition intervals of period T the end of the usefulinterval of period T following the said transition interval isretransmitted.

In a variant corresponding to the signals illustrated in FIG. 6c, theserialization device 301 supplies "O"s during the period ΔT of thetransition intervals.

The serialization device comprises means of storage and multiplexors.

It must be understood that other variants of generation of the signallike, for example the generation of homodyne signal using for example aplurality of devices for calculation of the Fourier transform does notdepart from the scope of the present invention.

In FIG. 12 can be seen an embodiment of the device 75 for conversion ofdigital words into complex signals. The device 75 comprises twoconversion tables 750 stored in devices with permanent memorycapability. For example, permanent memories of read-only memory,programmable read-only memory, erasable programmable read-only memory,electrically erasable programmable read-only memory or safeguardedrandom-access memory (ROM, PROM, EPROM, EEPROM or RAM) type are used.The digital words to be converted correspond to the addresses in thetables 750, the value of the amplitude or the real part of the signalbeing stored at this address in a first table 750, the value of thephase or the imaginary part of the signal being stored in the secondtable.

It must be understood that the two tables do not necessarily correspondto two memory boxes. Thus, it is possible to use a single memory boxhaving a sufficient capacity, or to use more than two memory boxes,depending on the desired resolution and the capacity of the memorycircuits used.

In FIG. 13 can be seen a rearrangement device 78. The rearrangementdevice 78 comprises a device with memory capability 781, a multiplexor782 and a sequencer 784. The device with memory capability 781 isconnected to the multiplexor 782. The sequencer 784 is connected, via acontrol line 785, to the device with memory capability 781 and via acontrol line 786, to the multiplexor 782. The rearrangement device 78enables the putting of the data to be processed into a format compatiblewith the calculating device 190. The rearrangement of the data depends,in particular, on the model, for example, for the circuits used forcalculation of the fast Fourier transform. The sequencer 784 enablesrearrangement of the order of digital words and/or bits inside digitalwords to be processed by the calculation circuit, not shown in FIG. 13.The sequencer 784 supplies the addresses to the device with memorycapability 781 by way of the line 785 as well as control signals. Thesequencer 784 supplies the control signals to the multiplexor 782 by wayof the line 786 enabling the switching between various positions of themultiplexor. The multiplexor 782 is, for example, a three-positionmultiplexor enabling the choosing between two memory banks and a zerogenerator 787. The zero generator 787 for example enables generation ofzeros necessary for the generation of the signal by the inverse Fouriertransform.

The zeros necessary for the generation of the inverse Fourier transformare stored in a device with memory capability 781. They are transmitted,either from special connections of the device with memory capability 781which is connected to the multiplexor 782, or by the addressing executedby the sequencer 784 of the device with memory capability 781.

If necessary the device 78 comprises an interface 783 enabling theadaptation of the output signals to the input signals of the circuit forcalculation, for example, of the discrete Fourier transform.

In FIG. 16 can be seen an embodiment of the device with memorycapability 781 of FIG. 13. In the example illustrated in FIG. 16 thedevice with memory capability 781 comprises four memory banks 7811,7812, 7813 and 7814. Each bank receives, for example, from the sequencer784 a read or write command R/W. Two banks, for example 7811 and 7812,are in read mode and two banks, for example 7813 and 7814, are in writephase simultaneously. Thus, the signals arriving are capable of beingwritten into a bank in the order which will be necessary for theirre-reading. The simultaneous re-reading of the second memory bankenables the supplying of the necessary digital data to the calculationcircuit.

In FIG. 17 can be seen a second embodiment of the device with memorycapability 781. The device with memory capability 781 in FIG. 17comprises only two memory banks 7811 and 7812. The sequencer 784 is, inthis case, a direct memory access (DMA) sequencer. Thus, the two memorybanks enable the simultaneous reading and writing of the data.

In the case of FIGS. 16 and 17 real and imaginary component I and Qphase quadrature data are simultaneously supplied.

In the device according to the present invention, it is possible to usethe modulation of the signal on various frequencies. For example, in thecase of high frequency electromagnetic wave usage it is possible tomodulate the signal directly on the transmission carrier, that is to sayat the transmission frequency, as illustrated in FIG. 15, to carry outthe modulation on intermediate frequencies as illustrated in FIG. 14 orto carry out the modulation on the base frequency.

The modulation on the baseband is necessarily carried out in terms of Iand Q. By contrast, at intermediate frequency or on the transmissioncarrier the modulation can be carried out from real signals asillustrated in FIG. 20.

The device of FIG. 20 comprises an analog-to-digital converter 3211, alow-pass filter 3209, a mixer 3201, a filter 3022, a mixer 3204 and afilter 3205 which are connected in series. The second inputs of themixers 3201 and 3204 are connected to local oscillators which are notshown in the figure.

In FIG. 14 can be seen a second embodiment of a device 302 forgeneration of the signal to be transmitted.

The device 302 comprises a first mixer 3201 and a second mixer 3207which are connected to a summation device 3023. The output of thesummation device 3023 is connected to a first input of a third mixer3204.

The second input of the mixer 3207 is connected to the output of a localoscillator 3305 generating the intermediate frequency. The second inputof the mixer 3201 is connected to the output of the local oscillator3305 by way of a device 3208 inducing a phase shift of π/2. Thus, theelevation in the frequencies of the phase quadrature components I and Qis carried out, the signal being reconstructed by the summation device3023.

The second input of the mixer 3204 is connected to a local oscillator3306 whose frequency of oscillation is higher than that of the localoscillator 3305.

Advantageously, the two oscillators 3305 and 3306 are synchronized by asingle time base, not shown. The local oscillators 3305 and 3304 aresufficiently stable to allow a reliable calibration at the receivingend.

Advantageously, the time base is synchronized with the device forsampling the signal.

In a variant the device 302 is a digital device.

In the variant illustrated in FIG. 14 the device 302 is an analog devicethus, it comprises, at the input end, analog-to-digital converters 3211and 3212 The converters 3211 and 3212 are connected to the first inputsof the mixer respectively 3201, 3207. Low-pass filters respectively3209, 3210 are placed between the output of the analog-to-digitalconverters 3211 and 3212 and the inputs of the mixers 3201, 3207. Thefilters 3209 and 3210 are intended to eliminate the high frequencycomponents generated by the analog-to-digital converters 3211 and 3212.

At the output of the mixers 3201, 3207 and 3204 it is necessary toposition filters respectively 3022, 3206 and 3205 which are intended toselect the desired part of the spectrum present at the output of themixers.

In FIG. 15 can be seen a variant of the device 302 comprising a singlefrequency elevation stage. The device 302 of FIG. 15 comprises a firstmixer 3201 and a second mixer 3207. The outputs of the mixers 3201 and3207 are connected to the inputs of a summation device 3203 by way ofband-pass filters 3022 and 3206. In the analog example illustrated inFIG. 15 the first input of the mixers 3201 and 3207 are connected to theoutputs of the analog-to-digital converters 3211 and 3212 by way of thefilters 3209 and 3210.

The transmitter according to the present invention conveys codingsignals enabling, at the receiving end, the precise synchronization of atime base of the receiver with a time base of the transmitter. Thus, itis possible to achieve a good temporal and/or phase resolution.

In an embodiment illustrated in FIG. 19 an analog synchronization isused.

In the example illustrated in FIG. 19 a set 3000 of modulated signals istransmitted on N channels, the spectrum is substantially rectangularhaving a frequency width f 1 equal to B, the passband, and a height Amcorresponding to the mean amplitude A 2 of the signal inside the band B.Two frequencies f_(A) and f_(B) with an amplitude AM considerablygreater than Am are transmitted. For example, AM is greater than Am by12 db. Thus, at the receiving end, by knowing the frequencies f_(A) andf_(B) it will be possible to separate f_(A) and f_(B). Through theknowledge, on the one hand, of the frequencies f_(A) and f_(B) and, onthe other hand, of their difference at the receiving end, a frequencyreference is obtained from which a time reference can be extracted. Atthe receiving end the difference f_(A) -f_(B) is obtained, for exampleby making the frequencies f_(A) and f_(B) beat in a mixer.

In an embodiment of the device according to the present invention B isequal to 8 MHz and f_(A) is separated from f_(B) by 5 MHz.

In FIG. 21 can be seen a diagram of an embodiment of a receiver on thepresent invention. The embodiment illustrated in FIG. 21 comprises areceiving aerial 40, an amplifier 603, a mixer 41, a band-pass filter42, a variable gain amplifier 604, a mixer 4817, a low-pass filter 4818,an analogdigital converter 4819, a reorthogonalization device 482, ademodulation device 48, a local oscillator 250, an automatic gaincontrol device 605, a local oscillator 491, a servocontrol device 49, ananalysis circuit 601, a decision circuit 602, a processing device 45,and an exploitation device 46.

The aerial 40 is connected to the input of the amplifier 603. The outputof the amplifier 603 is connected to a first input of the mixer 41. Theoutput of the mixer 41 is connected to the input of the band-pass filter42. The output of the band-pass filter 42 is connected to the input ofthe amplifier 604. The output of the amplifier 604 is connected on theone hand, to a first input of the mixer 4817, and on the other hand, tothe input of the automatic gain control circuit 605. The output of theautomatic gains control 605 is connected to a gain command input of theamplifier 604. The output of the mixer 4817 is connected to the input ofthe low pass filter 4818. The output of the low-pass filter 4818 isconnected to the input of the analog digital converter 4819. The outputof the analog digital converter 4819 is connected to the input of thereorthogonalization device 482. The output of the reorthogonalizationdevice 482 is connected to the input of the demodulation device 48. Theoutput of the demodulation device 48 is connected, on the one hand, tothe input of the analysis circuit 601 and on the other hand, to theinput of the servocontrol device 49. The output of the analysis circuit601 is connected to the input of the decision device 602. The output ofthe decision device 602 is connected to the input of the informationprocessing device 45. The output of the information processing device 45is connected to the input of the exploitation device 46. A first outputof the servocontrol device 49 is connected to the analog digitalconverter 4819, to the reorthogonalization device 482, to thedemodulation device 48, to the analysis circuit 601 and to the decisiondevice 602. A second output of the servocontrol device 49 is connectedto the local oscillator 491. A third output of the servocontrol device49 is connected to the local oscillator 250.

The aerial 40 receives the high frequency signal coming from thetransmitter.

The amplifier 603 amplifies the signal captured by the aerial 40. Bybeating with a high frequency signal supplied by the local oscillator250, the mixer 41 lowers the frequency of the received signal.

The signal is filtered by a filter 42. The filter 42 enables eliminationof the signals extraneous to the signals which one wishes to be able toreceive. The filter 42 is advantageously a surface acoustic wave (SAW)filter.

The amplifier 604 carries out, under the control of the automatic gaincircuit 605, the amplification of the intermediate frequency signal. Theautomatic gain circuit captures the signal at the output of theamplifier 604. An integration over a sufficiently long period of timesupplies the mean value of the amplitude of the signal for thecalculation of a command signal for the amplifier 604 enablingoptimization of the reception.

The mixer 4817 carries out beating between a signal supplied by thelocal oscillator 491 and the signals amplified by the amplifier 604. Themixer 4817 delivers a signal at the low carrier level. The filter 4818selects the desired part of the spectrum.

The analog-to-digital converter 4819 carries out the digital sampling ofthe signal.

In order to be able to obtain large information throughputs, it isparamount to carry out a total separation of the signals belonging tovarious channels. The reorthogonalization circuit 482 advantageouslyenables elimination of the crosstalk between channels. The crosstalkcould for example be the result of multiple echoes which delay part ofthe signal. Such signals arrive at the receiver, in particular, duringthe receiving of the subsequent pattern. The reorthogonalization device482 comprises a pattern-modification detection circuit. For example, itcomprises means for subtracting the signal from a signal delayed by aperiod T. While the two samples are taken in a single transmissioninterval of period T+ΔT their difference is almost constant. This istrue for each transmission interval during a period ΔT decreased by thearrival time of the most distant multiple echo. In contrast, the rapidfluctuation in this difference indicates that the two samples no longerbelong to the same transmission interval. Thus, from the difference intwo samples the instant of the transmission interval modification andconsequently a synchronization of the transmission intervals (alsocalled packet synchronization) is determined. The signal coming from themultiple echoes, being in danger of provoking a crosstalk, is eithereliminated in the case illustrated in FIG. 6a, or added in coherentfashion to the previous pattern in the case illustrated in FIG. 6c. Inthe first case, the period ΔT of the transition interval isadvantageously greater than the propagation period of the multipleechoes which it is desired to be able to eliminate. The elimination ofthe multiple echoes is carried out, for example, by not taking accountof the signals received during the transition intervals 81 of period ΔT.

In the second case, the signals arriving during the transition intervalare picked up and added to the start of the corresponding usefultransmission interval. This latter embodiment requires delay meansenabling storage of previous patterns prior to their processing by thedemodulation device 48.

The demodulation device 48 carries out the separation of the signalsbelonging to the various channels. In the example illustrated in thefigure, the processing is digital. For example, a device for calculationof the discrete Fourier transform is used. Advantageously, a device forcalculation of the fast Fourier transform (FFT) is used. However, ananalog separation for example by using frequency mixer banks, separatedby 1/T, does not depart from the scope of the present invention.

The demodulated signals are supplied, on the one hand, to an analysiscircuit 601, and on the other hand, to a servocontrol device 49.

The analysis circuit 601 carries out the analysis of the receivedsignals, the equalization and the calibration of the signals, fromcalibration or test signals received from the transmitter.

The servocontrol device 49 carries out the synchronization between thevarious stages of the receiver and between the receiver and thetransmitter. In particular, it supplies synchronization signals to thelocal oscillators 250 and 491 enabling their stable operation over time.Furthermore, it supplies a sampling frequency to the analog-to-digitalconverter 4819, to the reorthogonalization device 482, to thedemodulation device 48, to the analysis circuit 601 and to the decisiondevice 602.

The signals normalized by the analysis circuits 601 are supplied to thedecision device 602.

The decision circuit 602 determines which point 14 of FIG. 7 is involvedand hence which (real part of the signal, imaginary part of the signal)or (amplitude, phase) pair is involved. The decision device 602advantageously associates a digital word with each pair.

The receiver according to the present invention comprises other devicessuch as for example a processing device 45. The processing device 45carries out the desired processing on the signal. For example, in atelevision receiver the processing circuit 45 reconstructs the image andthe sound from digital signals. Advantageously, the processing device 45uses image decompression algorithms insofar as image compressionalgorithms were used at the transmission end.

The information processing device 45 is connected to the exploitationdevice 46. The exploitation device 46 enables exploitation of thesignals received. The type of exploitation device depends principally onthe type of receiver which is used. For example, for television signaltransmission a cathode ray tube or a flat screen and a loud speaker willin particular be used. For the transmission of telephonic data theexploitation device is, for example, a telephone exchange or atelephone. For the transmission of data the exploitation device 46 canbe, for example, a computer receiving the data to be processed or to bestored.

In FIG. 9 can be seen an embodiment of the receiver according to thepresent invention comprising, for the low carriers, a processing chainfor the real part and a processing chain for the imaginary part of thesignal in phase quadrature.

The device of FIG. 22 comprises an aerial 40, an amplifier 603, a mixer41, band-pass filter 42, a variable gain amplifier 604, an automaticgain command circuit 605, a mixer 4817, a mixer 4814, a low-pass filter4818, a low-pass filter 4815, an analog digital converter 4819, ananalog digital converter 4816, a reorthogonalization device 4821, areorthogonalization device 4822, a demodulation device 48 an analysiscircuit 601, a decision circuit 602, an information processing device45, a display device 462, a sound recording device 461, a servocontroldevice 49 and a τ/2 phase shifter 4813.

The aerial 40 is connected to the input of the amplifier 603. The outputof the amplifier 603 is connected to a first input of the mixer 41. Theoutput of the mixer 41 is connected to a band-pass filter 42. The outputof the band-pass filter 42 is connected to the input of an amplifier604. The output of the amplifier 604 is connected to the input of anautomatic gain command device 605, to a first input of a mixer 4817 andto a first input of the mixer 4814. The output of the automatic gaincommand device 605 is connected to a first gain command input of anamplifier 604. The output of the mixer 4817 is connected to the input ofthe filter 4818. The output of the mixer 4814 is connected to the inputof the filter 4815. The output of the filter 4818 is connected to theinput of the analog digital converter 4819. The output of the low-passfilter 4815 is connected to the input of the analog-to-digital converter4816. The output of the analog-to-digital converter 4819 is connected tothe input of the reorthogonalization device 4821. The output of theanalog-to-digital converter 4816 is connected to the input of thereorthogonalization device 4822. The outputs of the reorthogonalizationdevice 4821, 4822 are connected to the inputs of the demodulation device48. The output of the demodulation device 48 is connected to the inputof the analysis circuit 601 and of the servocontrol device 49. Theoutput of the analysis circuit 601 is connected to the input of thedecision device 602. The output of the decision device 602 is connectedto the input of the information processing device 45. The output of theinformation processing device 45 is connected to the exploitation devicesuch as for example, the display device 462 and sound recording device461. A first output of the servocontrol device 49 is connected to theanalog-to-digital converters 4819 and 4816, to the reorthogonalizationdevices 4821 and 4822, to the demodulation device 48, to the analysiscircuit 601 and to the decision device 602. This output supplies thesampling frequency. In the embodiment illustrated in FIG. 22 the beatfrequency is supplied directly via outputs 11 of the servocontrol device49. A high frequency output is connected to the second input of themixer 41. A middle frequency output is connected to the input of the π/2phase shifter 4813 and to the second input of the mixer 4817. The outputof the phase shifter 4813 is connected to the second input of the mixer4814. In the device illustrated in FIG. 22 the real and imaginary partof the signal, in phase quadrature, is operated on. Thus it is possibleto lower the frequency without losing information.

The demodulation device 48 comprises, advantageously, a device forcalculation of the Fourier transform.

Advantageously, the device for calculation of the Fourier transform is adevice for calculation of the discrete Fourier transform.

Advantageously, the device for calculation of the Fourier transform is adevice for calculation of the fast Fourier transform (FFT). The use offast Fourier transform algorithm necessitates the carrying out of thecalculations on a number of samples equal to a power of two. In thecourse of television transmissions, 256, 512, 1024 or 2048 channels arefor example used. However, it is not necessary that each channeltransmit an item of information. The use of a device for calculation ofthe fast Fourier transform in order to carry out the demodulation of thereceived signal enables use of standard circuits or a combination ofstandard circuits for calculation of the fast Fourier transform. It mustbe understood that other variants, such as for example homodynedemodulation, do not depart from the scope of the present invention.

In FIG. 23 can be seen an embodiment of the analysis device 601. Thedevice of FIG. 23 comprises a splitting device 586, equalization device587, a device for analyzing test signals 588 and a sequencer 585.

The splitting device 586 receives the signals to be processed. Theoutputs of the splitting device 586 are connected, on the one hand, tothe equalization device 587 and on the other hand, to the analysis andtest device 588. The output of the analysis and test device 588 isconnected, on the one hand, to the equalization device 587 and on theother hand, to the synchronization device 490.

The splitting device 586 separates the test signals which it conveystowards the analysis and test device 588, from the information signalswhich it conveys into the equalization device 587. The detection of thetest signals can be effected, for example, according to a specifiedtransmission standard. For example, the splitting device 586 "knows"that, in each transmission interval, a channel at the 8 level isreserved for the test signals. In another transmission standard, thetest signals can correspond to all the channels of a transmissioninterval at the, for example, 100 level. These two types of test signalsserving in the calibration of the received phase and/or amplitude can bemixed in order to give, for example, a test channel at the 16 levelevery 64 intervals.

In a first embodiment, the receiver according to the present inventionis designed to be able to follow a single standard. In such a case it isnecessary to carry out a first synchronization or indeed receive asynchronization from another device of the receiver.

In a second variant of the receiver according to the present inventionthe receiver can receive several transmission standards. In this case,it is necessary to detect which transition standard the received signalsbelong to. Insofar as the transmissions on separate channels enable thetransmission by multiplexing several channels and/or via a time-divisionmultiplexing to convey information of differing nature, it is possibleto reserve, for example, part of the information throughput for serviceinformation. The service information can, for example, containperiodically the item of information concerning the type of transmissioncarried out.

The transmission standard can also be chosen by switching by the userchoosing the desired programme. The latter chooses, for example, to passfrom a television transmission to a radiophonic transmission. Theinformation on the transmission standards is for example stored in apermament memory (not shown).

The splitting device 586 comprises, for example, multiplexors and ahardwired logic element carrying out the orders supplied by thesequencer 585.

The values of the test signals must be known to the receiver. Forexample, the test signals are pseudorandom signals. The signals aregenerated in the transmitter and the receiver according to the samealgorithm, which thus enables comparison of the received signal with asignal identical with the signal which was transmitted.

The analysis and test device 588 detects the level received in each ofthe test channels. It determines the phase shift and the attenuationwhich are received in the test channels. From the attenuation and thephase shift the analysis and test device 588 determines the attenuationsand the phase shifts in the channels intermediate between the testchannels by using, for example, the interpolation method. Theinterpolation can, for example, be a linear interpolation.

In order to obtain a large information throughput it is necessary to beable to distinguish close phases and amplitudes and hence to make use ofan amplitude and phase reference for each channel. This amplitude andphase reference is advantageously given by reference signals transmittedperiodically by the transmitter towards the receiver. The frequency ofrepetition of reference signals depends on the stability of thepropagation conditions and the local oscillators.

In a first variant of the device according to the present invention,phase and amplitude reference signals are periodically transmitted onall the frequencies 31 to 3N of a time interval of period T or T+ΔT.However, it is necessary to note that the frequent transmission ofcalibration signals reduces the throughput of useful transmittedinformation.

In an advantageous variant of the device according to the presentinvention only a few calibration signals are transmitted regularlydistributed advantageously over the frequencies 31 to 3 N, thecoefficients of the other frequencies being determined by calculation,for example by interpolation.

More generally, it is possible to distribute the calibration signals intime and/or on different channels.

It is for example possible to periodically transmit test signals, eachtransmission being made on different channels. For example, a circularpermutation of the channels assigned to the tests is carried out. Thepulsed response of the transmission medium is deduced for all thechannels, for example, by interpolation in time and/or on thefrequencies. The matrix of the corrections in amplitude and in phase tobe applied to each channel is thus deduced.

It is paramount to compensate, by calibrations, the variations in thepulsed response of the transmission medium, for example owing to avariation (even local) in the atmospheric conditions.

The pulsed response of the medium is determined, for example, bycalculating the Fourier transform of the corrections to be applied.

In one embodiment a channel at the 8 level served in the calibration ofthe amplitude A and of the phase of all the channels 31 to 3N. In such atype of device it is possible, either to carry out the calibration ateach useful transmission interval of period T, or as in the case of apreviously described embodiment, to solely reserve certain transmissionintervals for the calibration. The synchronization is maintained by theuse of a stable time base.

The analysis and test device comprises, for example, the device withmemory capability and microprocessors for processing of the rapidsignal. The values of phase shift and attenuations for each channel aretransmitted to the equalization device 587.

The equalization device 587 applies, to each channel, an amplificationand a phase shift which are inverse to those induced by thetransmission. Thus, the amplitudes of all the channels at the receivingend are, after equalization by the circuit 587, proportional to theamplitudes at the time of the transmission. In the same way, therelative phase shift between channels at the receiving end is, after theprocessing by the equalization device 587, equal to the relative phaseshift between the channels at the transmission end.

In an analog variant the equalization device 587 comprises variablephase shifters and variable amplifiers. The analog phase shifters canhave a digital control, charge transfer devices (CCD) comprising asingle input and a plurality of output can for example be used. Eachoutput corresponds to a different phase shift.

In a digital variant of the equalization device 587 multiplications andadditions are used to carry out amplitude and phase corrections.Hardwired logic elements and/or microprogrammed or programmed logicelements are used.

In FIG. 24 can be seen an architecture of known type capable of beingimplemented in the device according to the present invention. Thearchitecture of FIG. 24 is capable of being used in thereorthogonalization device. The device with memory capability 4841 is,for example, a two-port device. The data to be stored arrive via theinput port. These data, rearranged, set off again from the output port.The sequencer 4842 supplies the addresses for the inscription andrereading of the data. According to the desired type of datareorganization it is possible to reread entire words or only parts ofwords or individual bits. The device with memory capability 48-41comprises, for example, random-access memory (RAM) integrated circuits.

The sequencer 4842 comprises, for example, a hardwired logic element andcounters. In a variant, in order to use standard circuits, it ispossible to replace the sequencer 4842, for example, by amicroprocessor. Advantageously, the microprocessor is of the typeprocessing of the signal.

In FIG. 25 can be seen a second embodiment of a reorthogonalizationdevice. In the embodiment illustrated in FIG. 25, the device 482comprises a device with memory capability 4825, an arithmetic and logicunit 4826, a multiplexor 4823 as well as a sequencer 4824 The output ofthe device with memory capability 4825 is connected to the input of thearithmetic and logic unit 4826 and to a first input of the multiplexor4823. The output of the arithmetic and logic unit 4825 is connected to asecond input of the multiplexor 4823. The sequencer 4824 receives thegeneral signals for synchronization of the receiver, for example, from aservocontrol device 49. The sequencer 4824 sends control andsynchronization signals to the multiplexor 4823. The sequencer 4824sends address and synchronization signals to the device with memorycapability 4825. In the device illustrated in FIG. 25 the addressing ofthe memory 4825 by the sequencer 4824 enables the carrying out of therearrangement of the digital words. The arithmetic and logic unit ischarged with carrying out the desired summations of the signals. Theswitching of the multiplexor 4823 enables choice between the two modesof rearrangement depending on the desired transmission standard and thereceiving phase in progress.

Moreover, it is possible to use, in order to exploit the signal, to usean amplitude/phase demodulator such as that illustrated in FIG. 22 ofthe Patent Fr 86 139 37 filed by the Applicant on Oct. 7, 1986.

The transmitter according to the present invention conveys codingsignals enabling, at the receiving end, the precise synchronization of atime base of the receiver with a time base of the transmitter. Thus, itis possible to achieve a good temporal and/or phase resolution.

In a first embodiment of the device according to the present invention,a digital synchronization is used.

In an embodiment illustrated in FIG. 27, an analog synchronization isused.

In the example illustrated in FIG. 26, a set 3000 of modulated signalsis transmitted on N channels.

The spectrum is substantially rectangular having a frequency width f 1equal to B, the passband, and a height Am corresponding to the meanamplitude A 2 of the signal. Inside the band B, two frequencies f_(A)and f_(B) with an-amplitude AM considerably greater than Am aretransmitted. For example, AM is greater than Am by 12 db. Thus, at thereceiving end, by knowing the frequencies f_(A) and f_(B) it will bepossible to separate f_(A) and f_(B). Through the knowledge, on the onehand, of the frequency f_(A) and f_(B) and, on the other hand, of theirdifference at the receiving end, a reference of the frequency isobtained from which a time reference can be extracted. At the receivingend the difference f_(A) -F_(B) is obtained, for example by making thefrequencies f_(A) and f_(B) beat in a mixer.

In an embodiment of the device according to the present invention B isequal to 8 MHz and f_(A) is separated from f_(B) by 5 MHz.

In FIG. 14 is shown an analog embodiment of the servocontrol device 49of FIGS. 8 and 9. The device of FIG. 14 is intended to operate with asignal transmitted by the transmitter such as illustrated in FIG. 13.The servocontrol device 49 comprises a band-pass filter 701 and aband-pass filter 702, a mixer 703, a phase lock loop 704 (PLL), afrequency division phase lock loop 709 (PLL) , a frequency divisionphase lock loop 710 (PLL) and a frequency division phase lock loop 711(PLL). The phase lock loops comprise, for example, a mixer, a low-passfilter, a voltage controlled oscillator. In FIG. 27, the loop 704comprises a mixer 705, a low-pass filter 706, a voltage controlledoscillator 707 (VCO).

The input of the device 49 is connected to the inputs of the filters 701and 702. The output of the filter 701 is connected to a first input ofthe mixer 703. The output of the filter 702 is connected to a secondinput of the mixer 703. The output of the mixer 703 is connected to afirst input of the mixer 705. The output of the mixer 705 is connectedto the input of the low-pass filter 706. The output of the oscillator707 is connected to the input of the phase lock loop 709, to the inputof the phase lock loop 710 to the input of the phase lock loop 711. Theoutput of the low-pass filter 706 is connected to the input of theoscillator 707. The output of the oscillator 707 is connected to thesecond input of the mixer 705. The outputs of the phase lock loops 709,710 and 711 constitutes the outputs of the servocontrol device 49supplying the desired frequencies.

The filter 701 selects the frequency f_(A), the filter 702 selects thefrequency f_(B). The mixer 703 carries out the beating between thefrequency f_(A) and f_(B).

The phase lock loop 704 supplies the value of the differences betweenfrequencies f_(A) and f_(B). The difference between the frequenciesf_(B), f_(a) at the transmission end, determined by the transmissionstandard, is known. The comparison at the receiving end enables thesupplying of a frequency and phase reference.

The phase lock loops 709, 710 and 711 enable the supplying of frequencyand phase references which are sufficiently stable for the operation ofthe device according to the present invention. For example, the loop709, 710, 711 enables the supplying of a frequency referencerespectively to the local oscillator 250 and to the local oscillator 491of FIG. 21 and a sampling clock signal to the digital devices of FIGS. 8or 9. The oscillation frequencies depend on the settings of theoscillators.

The invention is applied to analog and/or digital information receivingdevices, to communications between computers, to telephoniccommunications between exchanges, to telephonic communications betweenradiotelephones and communications stations, to radioelectriccommunications between terrestrial stations and satellites, tocommunications between satellites, to acoustic communications in airand/or in water, to the construction of local computer networks and tothe receiving of radiophonic and television transmissions.

The present invention relates to a novel type of modulation being ableto be applied to all transmissions or acquisitions of information. Theinvention is applied to the device using all types of waves, inparticular acoustic waves, and more particularly, electromagnetic waves.

The device according to the present invention is applied, in particular,to radiophonic and television transmissions, to analog or digitalinformation transmission devices, to communications between computers,to telephonic communications between exchanges, to telephoniccommunications between radiotelephones and communications stations, toradioelectric communications between terrestrial stations andsatellites, to communications between two satellites, to acousticcommunications in air and/or in water, to the construction of localcomputer networks, to sonars, to radars.

The invention is particularly well suited to high fidelity radiophonictransmissions and reception as well as to high definition television(HDTV), and/or to digital television.

We claim:
 1. Method for simultaneous transmission of modulated wavesusing a plurality of orthogonal frequencies, in which symbols aretransmitted for a period T+ΔT, two transmission frequencies beingseparated by 1/T, T being the useful transmission interval and ΔT beingthe transition interval, absorbing the non-stationarities due to thearrival of echoes, in which during each transmission interval of periodT+ΔT a (real part, imaginary part) or (amplitude, phase) pair istransmitted on each frequency, the (real part, imaginary part) or(amplitude, phase) pair being in one-to-one equivalence with informationto be transmitted, characterised in that the number of pairs possible isgreater than 4, in that the symbols transmitted are constitutedperiodically by reference signals enabling, at a receiving end,equalization of the transmission channel, and in that synchronisationsignals are transmitted enabling, at the receiving end, processing ofthe signal during the useful transmission intervals of period T so as torecover tie orthogonality of the channels corresponding to theorthogonal frequencies.
 2. Method according to claim 1, characterised inthat a ratio ΔT/T is less than or equal to 1/8.
 3. Method according toclaim 1 or 2, characterised in that a first frequency used fo is equalto k/T, k being a positive integer or zero.
 4. Method according to anyone of claims 1 or 2, characterised in that it comprises a step:ofdetermination of patterns for the useful transmission interval of periodT, of transmission of the pattern during a transmission interval ofperiod T and its coherent continuation by recopying of the end of theuseful interval of the signal during the transition interval of periodΔT.
 5. Method according to any one of claims 1 or 2, characterised inthat the transmission is stopped during the transition intervals. 6.Method according to any one of claims 1 or 2, characterised in thatduring each useful transmission interval of period T, a symbol istransmitted on all or part of the frequencies.
 7. A transmitter systemfor simultaneous transmission of modulating wave using a plurality oforthogonal frequencies in which symbols are transmitted for a periodT+ΔT, two transmission frequencies being separated by 1/T, T being theuseful transmission interval and ΔT being the transition interval and inwhich during each transmission interval of period T+ΔT a realpart-imaginary part pair or amplitude-phase pair is transmitted on eachfrequency, the real part-imaginary part pair or amplitude-phase pairbeing in one-to-one equivalence with the information to be transmittedsaid system comprising:a coding device for receiving said information tobe transmitted, said coding device including a shaping means to producea desired shape of signals and a digital conversion means receiving theoutput of said shaping device and providing said real part-imaginarypart pair or amplitude-phase pair; and a modulation device for enablingthe transmission, during a useful transmission interval of said periodT+ΔT for each frequency used, of either a symbol chosen from a largeplurality of amplitude, phase pairs, or a reference signal to ensureequalization of the transmission channel.
 8. Transmitter according toclaim 7, characterised in that the modulation device (90) comprises adevice (190) for calculation of digital inverse Fourier transform formore than 1024 samples involving a processing time≦100 μs. 9.Transmitter according to either one of claims 7 and 8, wherein a channelis centered on the zero frequency carrier.
 10. Transmitter according toeither one of claims 7 and 8, characterised in that the modulationdevice (90) operates at intermediate frequency.
 11. Transmitteraccording to either one of claims 7 and 8, characterised in that themodulation device (90) is a digital device for carrier modulation.
 12. Areceiver system for receiving modulated waves transmitted with aplurality of orthogonal frequencies, in which symbols are transmittedfor a period T+ΔT, two transmission frequencies being separated by 1/T,T being the useful transmission interval and ΔT being the transmissioninterval in which, during each transition interval of said period, areal part-imaginary part pair or amplitude-phase pair is transmitted oneach frequency, the real part-imaginary part pair or amplitude-phasepair being in one-to-one equivalence with information to be transmitted,said receiver system comprising:means for transposing and sampling whichmeans are synchronous with a signal and wherein said means fortransposing and sampling include a means for demodulation of a modulatedwave transmission using symbols transmitted during said period T+ΔT onsaid plurality of orthogonal frequencies, said means for transposing andsampling further including a servocontrol device ensuring thesynchronization of the receiver system with a receive signal using saidtransition interval ΔT, said means for transposing and sampling furtherincluding a test means using reference signals for equalizingtransmission channels.
 13. Receiver according to claim 12, characterisedin that it comprises an automatic gain control device (AGC) controlledby a device for detecting the mean power of at least part of the signal.14. Receiver according to either one of claims 12 and 13, characterisedin that it comprises at least one device (48) for calculation of thefast Fourier transform (FFT) for more than 1024 samples involving aprocessing time ≦100 μs.
 15. Receiver according to claim 14,characterised in that it comprises (real part, imaginary part) or(amplitude, phase) pair decoding means (45), in order to convert theminto digital words.
 16. Receiver according to claim 12, characterised inthat the test means comprise a device (587) for equalisation,compensating for the perturbations in the signal coming from thetransmission and in particular multiple paths due to said echoes. 17.Receiver according to any one of claims 12, 13, 15, or 16, characterisedin that it comprises reorthogonalisation means (482, 4821, 4822) usingthe transition interval ΔT in order to render a plurality of channelsorthogonal.
 18. Receiver according to claim 17, characterised in thatthe reorthogonalisation means comprise a pattern change detectioncircuit.
 19. Receiver according to claim 18, characterised in that thepattern change detection circuit comprises means for subtraction of thesignal with a signal delayed by a period T and means for determiningwhether the difference is almost constant or not.